1. Field of the Invention
The present inventions relate to directional couplers and terminations for coaxial and stripline conductors.
2. Description of the Related Art
Directional couplers are useful measurement tools which provide a simple, convenient, accurate means for sampling microwave energy. Directional couplers also provide the ability to separate forward from reflected power.
FIG. 1 illustrates the basic construction of a conventional coupled-line directional coupler 10 useful in, for example, microwave applications. The directional coupler 10 consists of first and second parallel striplines 11, 12 coupled over multiples of approximately one-quarter wavelength (.lambda./4). Ports A and B are connected to first stripline 11 and port D and port C are connected to the second stripline 12.
The first and second parallel striplines 11, 12, referred to respectively as main and auxiliary lines, are separated from each other except in a coupling region 26. Ports A-C are usually configured for connection to coaxial transmission lines and the outer conductor or ground for each coaxial line is connected to grounded body 22 of coupler 10. Port D 18 terminates the second stripline 12 by interconnecting stripline 12 to the body 22 of coupler 10, which is at ground potential, through resistor 24.
In the coupling region 26, energy applied to main line 11 is directionally coupled from the main line 11 to the auxiliary line 12 and vice versa. In particular, energy applied at port A of the main line 10 appears at port B of main line 11; however, some fraction of the energy will appear at port C of the auxiliary line 12. The amount of energy appearing at port C of auxiliary line 12 depends upon the amount of coupling provided in the design of the unit. Several factors, including the spacing between lines 11, 12, determine the amount of energy that may be transferred from the one line 11, 12 to the other. The amount of coupling desired for forward power--power flowing in the port A-to-port-B direction--varies with the application. For example, a coupler used to split a signal would use a large amount of coupling. Coupling values from 3 dB to beyond 30 dB are typically encountered in practice.
Energy applied to port B of main line 10 will appear at port A, but practically none of this energy will appear at port C. The degree of discrimination in auxiliary line 12 between energy flowing in the port B-to-port-A direction and energy flowing in the port A-to-port-B direction is the directivity of the coupler. Directivity is calculated as the ratio of the forward-to-reverse coupling, expressed in dB, and is a measure of isolation obtainable at coupled port C with power being fed into the main line 11 at port B. The intention is to ensure that a minimum of the energy flowing in the port B-to-port-A direction will reach a load connected to port C of the auxiliary line 12, and thus the ideal directional coupler will have an infinite value of directivity. Values of directivity are usually low, on the order of 5 to 30 dB.
A directional coupler is also a useful device for measuring reflected energy. This is accomplished by applying energy to port B and connecting a device under test at port A. Energy reflected by the device under test will flow in the port A-to-port-B direction and a known fraction thereof will appear at port C.
All parallel-line couplers, whether true TEM or quasi-TEM, have an odd mode and an even mode property which results in odd and even mode impedances Z.sub.Oo and Z.sub.Oe. Directional couplers operating in the true TEM mode ideally yield equal phase velocities for the odd and even modes; however, most true TEM mode directional couplers, as well as quasi-TEM transmission lines (for example, microstrips) and other structures, yield different odd mode and even mode phase velocities, v.sub.po and v.sub.pe.
The propagation velocity (or phase velocity) v.sub.p of a wave traveling along a transmission line is ##EQU1## where L is the inductance of the transmission line, C is the capacitance of the transmission line, .mu..sub.r and .epsilon..sub.r are the permeability and permittivity of the medium through which the wave passes, and c is the velocity of light in free space. Most transmission lines do not comprise any ferromagnetic materials, and thus .mu..sub.r =1. Accordingly, for a uniform dielectric surrounding a conductor ##EQU2##
Coupled striplines 11, 12 of conventional directional couplers are constructed using metalized plastic layers similar to multi-layer printed circuit board. The metal is etched to form a desired conductor or circuit pattern. Very small, high-frequency geometries can be formed this way; however, the radio frequency (RF) performance is never very good because of three major problem areas: First, the dimensional tolerance (particularly the thickness) of the plastic layers is large; second, it is difficult to provide connections from the stripline structure to coaxial connectors with little RF reflection, especially at the terminated port, port D 18 which is largely responsible for the directivity of the coupler; and third, small air gaps between the layers of plastic contribute to differing propagation velocities of the odd and even TEM modes. In the case of conductors formed of metalized plastic layers, the non-uniformity of the dielectric causes the transmission lines to have differing odd and even mode propagation velocities, i.e., v.sub.po .noteq.v.sub.pe. The difference in the propagation or phase velocities of the odd and even modes degrades directivity. Differing propagation velocities of the odd and even modes have several causes, and thus result even if the dielectric medium which supports the striplines 11, 12 is uniform.
For parallel-coupled microstrips a dielectric overlay of substrate-type dielectric material can be provided in the region over the coupled microstrip lines. This dielectric overlay is useful for reducing the odd-mode phase velocity. See, T. C. Edwards, Foundations of Microstrip Design, John Wiley & Sons, p. 151. However, microstrip couplers, even those utilizing dielectric overlays, do not provide satisfactory results. In particular, the propagation velocities v.sub.pe and v.sub.po are dependent on frequency, and the geometry of the microstrips, causing a phenomenon known as dispersion. These problems relate to the fact that with microstrips, including those having dielectric overlays, the electric fields pass from air to another dielectric. Accordingly, the propagation velocity is dependent on some combination or average of .epsilon..sub.0 and .epsilon..sub.r, where .epsilon..sub.r is the permittivity of the dielectric.
Another type of construction for the striplines 11, 12 is a relatively thick self-supporting metal where the air surrounding the striplines is the dielectric. The shape of a self-supporting metal stripline is usually provided by machining the stripline in a step-like fashion (see FIG. 6) providing so-called stepped striplines. These steps in the striplines correspond to quarter wavelength sections and are another factor which causes the odd and even modes to have different propagation velocities. Although it is possible to manufacture self-supporting metal striplines with a smooth taper as opposed to steps, the design and manufacture of such striplines is very difficult.
The designation, as used herein, stripline refers to any conductor which has infinite ground planes on both sides of the conductor. The conductor itself may have different shapes, e.g., round or rectangular. This structure is difficult to construct for high-frequency applications because of the necessary small size of the striplines and tight tolerances. Fabricating a round conductor having quarter wavelength sections is particularly difficult Indeed, until 1984, it was believed that the maximum frequency which could be handled by coaxial couplers was 26 GHz. This perceived limitation was related, at least in part, to the inability to manufacture components such as couplers with the small dimensions and tolerances required for frequencies above 26 GHz. Tolerances on the order of approximately 0.0005" must be maintained for frequencies over 26 GHz.
Port D 18 includes a termination which ideally absorbs all of the RF energy impinged thereon. FIG. 2 shows a conventional termination for a coaxial conductor. A conductor 28, which may be the centerline of a coaxial transmission line or a stripline in a directional coupler, is connected to a resistor 30 provided between conductor 28 and the outer conductor of a coaxial transmission line or other ground plane 32. The outer conductor or ground plane 32 must be precisely shaped to effectively transform from Z.sub.o (the coaxial or conductor impedance) to zero impedance at the ground end. In particular, the distance between the outer conductor 32 and resistor 30 and the rate of change in this distance is important. Conventional resistors 30 are rod-shaped and have a diameter approximately the same as the diameter of the coaxial conductor; such resistors 30 are fabricated by providing a resistive film 34 on a ceramic rod 36. However, it is difficult to provide precise ceramic rods 36--again dimensions and tolerances are important--and to provide a uniform and mechanically accurate resistive film 34.
The quality of a coaxial termination depends in large part on the following factors: (1) the DC value of the resistor and the variation of this value with time and temperature; (2) the RF design of each element in order to have a minimum reflection; and (3) the ability to have close mechanical tolerances during manufacturing--any deviation from the perfect RF design causes reflection.
Conventional "rod" resistors 30 present problems with respect to all three of these factors, particularly in designs for higher frequencies which necessitate small components. Further, it is difficult to connect such resistors to the centerline conductor of a coaxial transmission line. The small dimensions required for high-frequency RF applications make it nearly impossible to manufacture the outer conductor 32 in the required shape.
To avoid the problems associated with conventional rod-shaped resistors 30 and ground planes 32 having a circular cross-section, planar resistors have been used for coaxial terminations. However, with a planar resistor the conductor is not coaxial and the RF waves must undergo a mode change from coaxial to planar at the junction of the coaxial centerline and the planar resistor. Such mode changes have conventionally been abrupt (or hard) mode changes.
FIG. 3 illustrates the structure which causes a conventional hard mode change when a transmitted signal passes between a coaxial conductor having a centerline conductor 38 and a ground plane or outer conductor 40 and a planar conductor, for example, a planar resistor 42 having a substrate 44 with resistive films 46 provided on the substrate 44. A hard or abrupt mode change causes a fairly large reflection, thereby degrading the quality of the RF termination.
An additional problem with conventional directional couplers and terminations is that it is difficult to support metal striplines which utilize an air dielectric. This problem is compounded at port D 18 where the stripline 12 is connected to a termination. Conventionally, a support mechanism is provided between the stripline 12 and the ground body 22 of a coupler to support stripline 12. Such support mechanisms almost always cause reflections of the signals transmitted by stripline 12. In addition, support mechanisms require mode changes from stripline to the coaxial mode of the support mechanism and the termination. A conventional termination as shown in FIG. 2 is connected to the support mechanism.